AD8551

a

AD8551/AD8552/AD8554 Zero-Drift, Single-Supply,

Rail-to-Rail Input/Output Operational Amplifiers

8-Lead SOIC (R Suffix)

1 2 3 4

8 7 6 5

AD8551

2IN A V2 +IN A

V+

OUT A NC

NC NC

NC = NO CONNECT

1 2 3 4

8 7 6 5

AD8552

2IN A V2 +IN A

OUT B 2IN B V+

+IN B OUT A

14-Lead SOIC (R Suffix)

14 13 12 11 10 9 8 1

2 3 4 5 6 7 2IN A

+IN A V+

+IN B 2IN B OUT B

OUT D 2IN D +IN D V2 +IN C 2IN C OUT C OUT A

AD8554 FEATURES

Low Offset Voltage: 1 ␮V Input Offset Drift: 0.005 ␮V/ⴗC Rail-to-Rail Input and Output Swing +5 V/+2.7 V Single-Supply Operation High Gain, CMRR, PSRR: 130 dB Ultralow Input Bias Current: 20 pA Low Supply Current: 700 ␮A/Op Amp Overload Recovery Time: 50 ␮s No External Capacitors Required APPLICATIONS

Temperature Sensors Pressure Sensors

Precision Current Sensing Strain Gage Amplifiers Medical Instrumentation Thermocouple Amplifiers GENERAL DESCRIPTION

This new family of amplifiers has ultralow offset, drift and bias current. The AD8551, AD8552 and AD8554 are single, dual and quad amplifiers featuring rail-to-rail input and output swings. All are guaranteed to operate from +2.7 V to +5 V single supply.

The AD855x family provides the benefits previously found only in expensive autozeroing or chopper-stabilized amplifiers. Using Analog Devices’ new topology these new zero-drift amplifiers combine low cost with high accuracy. No external capacitors are required.

With an offset voltage of only 1 µV and drift of 0.005 µV/°C, the AD8551 is perfectly suited for applications where error sources cannot be tolerated. Temperature, position and pres- sure sensors, medical equipment and strain gage amplifiers benefit greatly from nearly zero drift over their operating temperature range. The rail-to-rail input and output swings provided by the AD855x family make both high-side and low- side sensing easy.

The AD855x family is specified for the extended industrial/

automotive (–40°C to +125°C) temperature range. The AD8551 single is available in 8-lead MSOP and narrow 8-lead SOIC packages. The AD8552 dual amplifier is available in 8-lead narrow SO and 8-lead TSSOP surface mount packages. The AD8554 quad is available in narrow 14-lead SOIC and 14-lead TSSOP packages.

8-Lead MSOP (RM Suffix)

2IN A 1IN A V2

V+

OUT A NC 1

4 5

8

AD8551

NC

NC = NO CONNECT NC

8-Lead TSSOP (RU Suffix)

2IN A +IN A V2

OUT B 2IN B +IN B V+

1

4 5

8

AD8552

OUT A

14-Lead TSSOP (RU Suffix)

OUT A 2IN A 1IN A V1

2IN D 1IN D V2 OUT D

1IN B 2IN B

OUT B 2IN C

OUT C 1IN C

AD8554

1 14

7 8

PIN CONFIGURATIONS

8-Lead SOIC (R Suffix)

ELECTRICAL CHARACTERISTICS

Parameter Symbol Conditions Min Typ Max Units

INPUT CHARACTERISTICS␣

Offset Voltage V_OS 1 5 µV

–40°C ≤ TA ≤ +125°C 10 µV

Input Bias Current I_B 10 50 pA

–40°C ≤ TA ≤ +125°C 1.0 1.5 nA

Input Offset Current I_OS 20 70 pA

–40°C ≤ TA ≤ +125°C 150 200 pA

Input Voltage Range 0 5 V

Common-Mode Rejection Ratio CMRR VCM = 0 V to +5 V 120 140 dB

–40°C ≤ T_A≤ +125°C 115 130 dB

Large Signal Voltage Gain¹ AVO RL = 10 kΩ, VO = +0.3 V to +4.7 V 125 145 dB

–40°C ≤ T_A≤ +125°C 120 135 dB

Offset Voltage Drift ∆VOS/∆T –40°C ≤ TA ≤ +125°C 0.005 0.04 µV/°C

OUTPUT CHARACTERISTICS

Output Voltage High VOH RL = 100 kΩ to GND 4.99 4.998 V

–40°C to +125°C 4.99 4.997 V

RL = 10 kΩ to GND 4.95 4.98 V

–40°C to +125°C 4.95 4.975 V

Output Voltage Low VOL RL = 100 kΩ to V+ 1 10 mV

–40°C to +125°C 2 10 mV

RL = 10 kΩ to V+ 10 30 mV

–40°C to +125°C 15 30 mV

Short Circuit Limit ISC ±25 ±50 mA

–40°C to +125°C ±40 mA

Output Current IO ±30 mA

–40°C to +125°C ±15 mA

POWER SUPPLY␣

Power Supply Rejection Ratio PSRR V_S = +2.7 V to +5.5 V 120 130 dB

–40°C ≤ T_A ≤ +125°C 115 130 dB

Supply Current/Amplifier I_SY V_O = 0 V 850 975 µA

–40°C ≤ T_A ≤ +125°C 1,000 1,075 µA

DYNAMIC PERFORMANCE␣

Slew Rate SR RL = 10 kΩ 0.4 V/µs

Overload Recovery Time 0.05 0.3 ms

Gain Bandwidth Product GBP 1.5 MHz

NOISE PERFORMANCE␣

Voltage Noise en p-p 0 Hz to 10 Hz 1.0 µV p-p

e_n p-p 0 Hz to 1 Hz 0.32 µV p-p

Voltage Noise Density en f = 1 kHz 42 nV/√Hz

Current Noise Density i_n f = 10 Hz 2 fA/√Hz

NOTE

1Gain testing is highly dependent upon test bandwidth.

Specifications subject to change without notice.

(V_S = +5 V, V_CM = +2.5 V, V_O = +2.5 V, T_A = +25ⴗC unless otherwise noted)

AD8551/AD8552/AD8554 ELECTRICAL CHARACTERISTICS

Parameter Symbol Conditions Min Typ Max Units

INPUT CHARACTERISTICS␣

Offset Voltage V_OS 1 5 µV

–40°C ≤ TA ≤ +125°C 10 µV

Input Bias Current I_B 10 50 pA

–40°C ≤ TA ≤ +125°C 1.0 1.5 nA

Input Offset Current I_OS 10 50 pA

–40°C ≤ TA ≤ +125°C 150 200 pA

Input Voltage Range 0 2.7 V

Common-Mode Rejection Ratio CMRR VCM = 0 V to +2.7 V 115 130 dB

–40°C ≤ T_A≤ +125°C 110 130 dB

Large Signal Voltage Gain¹ AVO RL = 10 kΩ, VO = +0.3 V to +2.4 V 110 140 dB

–40°C ≤ T_A≤ +125°C 105 130 dB

Offset Voltage Drift ∆VOS/∆T –40°C ≤ TA ≤ +125°C 0.005 0.04 µV/°C

OUTPUT CHARACTERISTICS

Output Voltage High VOH RL = 100 kΩ to GND 2.685 2.697 V

–40°C to +125°C 2.685 2.696 V

RL = 10 kΩ to GND 2.67 2.68 V

–40°C to +125°C 2.67 2.675 V

Output Voltage Low VOL RL = 100 kΩ to V+ 1 10 mV

–40°C to +125°C 2 10 mV

RL = 10 kΩ to V+ 10 20 mV

–40°C to +125°C 15 20 mV

Short Circuit Limit ISC ±10 ±15 mA

–40°C to +125°C ±10 mA

Output Current IO ±10 mA

–40°C to +125°C ±5 mA

POWER SUPPLY␣

Power Supply Rejection Ratio PSRR V_S = +2.7 V to +5.5 V 120 130 dB

–40°C ≤ T_A ≤ +125°C 115 130 dB

Supply Current/Amplifier I_SY V_O = 0 V 750 900 µA

–40°C ≤ T_A ≤ +125°C 950 1,000 µA

DYNAMIC PERFORMANCE␣

Slew Rate SR RL = 10 kΩ 0.5 V/µs

Overload Recovery Time 0.05 ms

Gain Bandwidth Product GBP 1 MHz

NOISE PERFORMANCE␣

Voltage Noise en p-p 0 Hz to 10 Hz 1.6 µV p-p

Voltage Noise Density e_n f = 1 kHz 75 nV/√Hz

Current Noise Density in f = 10 Hz 2 fA/√Hz

NOTE

1Gain testing is highly dependent upon test bandwidth.

Specifications subject to change without notice.

(V_S = +2.7 V, V_CM = +1.35 V, V_O = +1.35 V, T_A = +25ⴗC unless otherwise noted)

ORDERING GUIDE

Temperature Package Package

Model Range Description Option Brand¹

AD8551ARM² –40°C to +125°C 8-Lead MSOP RM-8 AHA

AD8551AR –40°C to +125°C 8-Lead SOIC SO-8

AD8552ARU³ –40°C to +125°C 8-Lead TSSOP RU-8

AD8552AR –40°C to +125°C 8-Lead SOIC SO-8

AD8554ARU³ –40°C to +125°C 14-Lead TSSOP RU-14

AD8554AR –40°C to +125°C 14-Lead SOIC SO-14

NOTES

1Due to package size limitations, these characters represent the part number.

2Available in reels only. 1,000 or 2,500 pieces per reel.

3Available in reels only. 2,500 pieces per reel.

ABSOLUTE MAXIMUM RATINGS

Supply Voltage . . . +6 V Input Voltage . . . GND to VS + 0.3 V Differential Input Voltage² . . . ±5.0 V ESD(Human Body Model) . . . 2,000 V Output Short-Circuit Duration to GND . . . Indefinite Storage Temperature Range

RM, RU and R Packages . . . –65°C to +150°C Operating Temperature Range

AD8551A/AD8552A/AD8554A . . . –40°C to +125°C Junction Temperature Range

RM, RU and R Packages . . . –65°C to +150°C Lead Temperature Range (Soldering, 60 sec) . . . +300°C

NOTES

1Stresses above those listed under Absolute Maximum Ratings may cause perma- nent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those listed in the operational sections of this specification is not implied. Exposure to absolute maximum rating condi- tions for extended periods may affect device reliability.

2Differential input voltage is limited to ±5.0 V or the supply voltage, whichever is less.

Package Type ␪^JA1 ␪^JC Units

8-Lead MSOP (RM) 190 44 °C/W

8-Lead TSSOP (RU) 240 43 °C/W

8-Lead SOIC (R) 158 43 °C/W

14-Lead TSSOP (RU) 180 36 °C/W

14-Lead SOIC (R) 120 36 °C/W

NOTE

1θJA is specified for worst case conditions, i.e., θJA is specified for device in socket for P-DIP packages, θJA is specified for device soldered in circuit board for SOIC and TSSOP packages.

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD8551/AD8552/AD8554 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.

WARNING!

ESD SENSITIVE DEVICE

AD8551/AD8552/AD8554

OFFSET VOLTAGE – mV

NUMBER OF AMPLIFIERS

180

0

22.5 0.5

120

100

60

20

2.5 V_SY = +2.7V V_CM = +1.35V T_A = +258C

40 80 140 160

21.5 20.5 1.5

Figure 1. Input Offset Voltage Distribution at +2.7 V

OFFSET VOLTAGE – mV

NUMBER OF AMPLIFIERS

180

0 120

100

60

20

V_SY = +5V V_CM = +2.5V T_A = +258C

40 80 140 160

22.5 21.5 20.5 0.5 1.5 2.5

Figure 4. Input Offset Voltage Distribution at +5 V

LOAD CURRENT – mA 10

0.1 0.001

OUTPUT VOLTAGE – mV

0.1 1 10

1 100 10k

SOURCE SINK V_SY = +2.7V

T_A = +258C

100 1k

0.0001 0.01

Figure 7. Output Voltage to Supply Rail vs. Output Current at +2.7 V

INPUT COMMON-MODE VOLTAGE – V

INPUT BIAS CURRENT – pA

50

230

0 1 2 3 4 5

40 30 20

210 220 10 0

V_SY = +5V

T_A = 2408C, +258C, +858C

2408C +258C +858C

Figure 2. Input Bias Current vs.

Common-Mode Voltage

INPUT OFFSET DRIFT – nV/8C

NUMBER OF AMPLIFIERS

12

0

0 1 2 3 4 5 6

10

8

4

2 6

V_SY = +5V V_CM = +2.5V T_A = 2408C TO +1258C

Figure 5. Input Offset Voltage Drift Distribution at +5 V

TEMPERATURE – 8C

INPUT BIAS CURRENT –pA

0

21000

275250225 0 25 50 75 100 125 2250

2500

2750

150 V_CM = +2.5V

V_SY = +5V

Figure 8. Bias Current vs. Temperature

INPUT COMMON-MODE VOLTAGE – V

INPUT BIAS CURRENT – pA

1,500

22,000

0 1 2 3 4 5

1,000

500

0

21,000 21,500 2500

V_SY = +5V T_A = +1258C

Figure 3. Input Bias Current vs.

Common-Mode Voltage

LOAD CURRENT – mA 10

0.1 0.001

OUTPUT VOLTAGE – mV

0.1 1 10

1 100 10k

100 1k

0.0001 0.01 SOURCE

SINK V_SY = +5V

T_A = +258C

Figure 6. Output Voltage to Supply Rail vs. Output Current at +5 V

TEMPERATURE – 8C

SUPPLY CURRENT –mA

1.0

0.8

0275250225 0 25 50 75 100 125 0.6

0.4

0.2

150 +5V

+2.7V

Figure 9. Supply Current vs.

Temperature

Typical Performance Characteristics–

SUPPLY VOLTAGE – V SUPPLY CURRENT PER AMPLIFIER – mA 800

0 700

400

300

200

100 600

500

0 1 2 3 4 5 6

T_A = +258C

Figure 10. Supply Current vs.

Supply Voltage

FREQUENCY – Hz

CLOSED-LOOP GAIN – dB

100 1k 10k 100k 1M 10M

60 50

240 40 30 20 10 0 210 220 230

A_V = 2100

V_SY = +2.7V C_L = 0pF R_L = 2kV

A_V = 210 A_V = +1

Figure 13. Closed Loop Gain vs.

Frequency at +2.7 V

V_SY = +5V

A_V = 100

A_V = 1 A_V = 10

FREQUENCY – Hz

OUTPUT IMPEDANCE – V

100 1k 10k 100k 1M 10M

300 270

0 240 210 180 150 120 90 60 30

Figure 16. Output Impedance vs.

Frequency at +5 V

FREQUENCY – Hz

OPEN-LOOP GAIN – dB

10k 100k 1M 10M 100M

60 50

240 40 30 20 10 0 210 220 230

45 90 135 180 225 270 0

PHASE SHIFT – Degrees

V_SY = +2.7V C_L = 0pF R_L =

Figure 11. Open-Loop Gain and Phase Shift vs. Frequency at +2.7 V

A_V = 2100

V_SY = +5V C_L = 0pF R_L = 2kV

A_V = 210 A_V = +1

FREQUENCY – Hz

CLOSED-LOOP GAIN – dB

100 1k 10k 100k 1M 10M

60 50

240 40 30 20 10 0 210 220 230

Figure 14. Closed Loop Gain vs.

Frequency at +5 V

2ms 500mV

V_SY = +2.7V C_L = 300pF R_L = 2kV A_V = +1

Figure 17. Large Signal Transient Response at +2.7 V

FREQUENCY – Hz

OPEN-LOOP GAIN – dB

10k 100k 1M 10M 100M

60 50

240 40 30 20 10 0 210 220 230

45 90 135 180 225 270 0

PHASE SHIFT – Degrees

V_SY = +5V C_L = 0pF R_L =

Figure 12. Open-Loop Gain and Phase Shift vs. Frequency at +5 V

FREQUENCY – Hz

OUTPUT IMPEDANCE – V

100 1k 10k 100k 1M 10M

300 270

0 240 210 180 150 120 90 60 30

V_SY = +2.7V

A_V = 100

A_V = 1 A_V = 10

Figure 15. Output Impedance vs.

Frequency at +2.7 V

5ms 1V

V_SY = +5V C_L = 300pF R_L = 2kV A_V = +1

Figure 18. Large Signal Transient Response at +5 V

AD8551/AD8552/AD8554

5ms 50mV

V_SY = 61.35V C_L = 50pF R_L = A_V = +1

Figure 19. Small Signal Transient Response at +2.7 V

CAPACITANCE – pF

SMALL SIGNAL OVERSHOOT – %

10 100 1k 10k

45

0 40 35 30 25 20 15 10 5

+OS 2OS V_SY = 62.5V

R_L = 2kV T_A = +258C

Figure 22. Small Signal Overshoot vs. Load Capacitance at +5 V

200ms 1V

VS = 62.5V RL = 2kV AV = 2100 VIN = 60mV p-p

Figure 25. No Phase Reversal

5ms 50mV

V_SY = 62.5V C_L = 50pF R_L = A_V = +1

Figure 20. Small Signal Transient Response at +5 V

V_SY = 62.5V V_IN = 2200mV p-p (RET TO GND) C_L = 0pF R_L = 10kV A_V = 2100

20ms 1V

0V V_IN

V_OUT

0V

BOTTOM SCALE: 1V/DIV TOP SCALE: 200mV/DIV

Figure 23. Positive Overvoltage Recovery

V_SY = +2.7V

FREQUENCY – Hz

CMRR – dB

140

80

0

100 1k 10k 100k 1M 10M

60 120

20 40 100

Figure 26. CMRR vs. Frequency at +2.7 V

CAPACITANCE – pF

SMALL SIGNAL OVERSHOOT – %

10 100 1k 10k

50 45

0 40 35 30 25 20 15 10 5

+OS 2OS V_SY = 61.35V

R_L = 2kV T_A = +258C

Figure 21. Small Signal Overshoot vs. Load Capacitance at +2.7 V

V_SY = 62.5V V_IN = +200mV p-p (RET TO GND) C_L = 0pF R_L = 10kV A_V = 2100

20ms 1V

V_IN 0V

0V

V_OUT

BOTTOM SCALE: 1V/DIV TOP SCALE: 200mV/DIV

Figure 24. Negative Overvoltage Recovery

FREQUENCY – Hz

CMRR – dB

140

80

0

100 1k 10k 100k 1M 10M

60 120

20 40 100

V_SY = +5V

Figure 27. CMRR vs. Frequency at +5 V

FREQUENCY – Hz

PSRR – dB

140

80

0

100 1k 10k 100k 1M 10M

60 120

20 40 100

+PSRR 2PSRR

V_SY = 61.35V

Figure 28. PSRR vs. Frequency at ±1.35 V

3.0 2.5 2.0 1.5

0.5 1.0

V_SY = 62.5V R_L = 2kV A_V = +1 THD+N < 1%

T_A = +258C 3.5

4.0 4.5 5.0 5.5

FREQUENCY – Hz

OUTPUT SWING – V p-p

0

100 1k 10k 100k 1M

Figure 31. Maximum Output Swing vs. Frequency at +5 V

en – nV/ Hz

V_SY = +2.7V R_S = 0V

0.5

FREQUENCY – kHz

1.0 1.5 2.0 2.5

0 52 78 104 130 156 182

26

Figure 34. Voltage Noise Density at +2.7 V from 0 Hz to 2.5 kHz

+PSRR 2PSRR

V_SY = 62.5V

FREQUENCY – Hz

PSRR – dB

140

80

0

100 1k 10k 100k 1M 10M

60 120

20 40 100

Figure 29. PSRR vs. Frequency at ±2.5 V

1s 2mV

V_SY = 61.35V A_V = 10,000

0V

Figure 32. 0.1 Hz to 10 Hz Noise at +2.7 V

en – nV/ Hz

V_SY = +2.7V R_S = 0V

5

FREQUENCY – kHz

10 15 20 25

0 32 48 64 80 96 112

16

Figure 35. Voltage Noise Density at +2.7 V from 0 Hz to 25 kHz

FREQUENCY – Hz

OUTPUT SWING – V p-p

3.0

2.5

0100 1k 10k 100k 1M

2.0

1.5

0.5 1.0

V_SY = 61.35V R_L = 2kV A_V = +1 THD+N < 1%

T_A = +258C

Figure 30. Maximum Output Swing vs. Frequency at +2.7 V

1s 2mV

V_SY = 62.5V A_V = 10,000

Figure 33. 0.1 Hz to 10 Hz Noise at +5 V

V_SY = +5V R_S = 0V

0.5

FREQUENCY – kHz

1.0 1.5 2.0 2.5

0 26 39 52 65 78 91

13 en – nV/ Hz

Figure 36. Voltage Noise Density at +5 V from 0 Hz to 2.5 kHz

AD8551/AD8552/AD8554

en – nV/ Hz

5 10 15 20 25

0

V_SY = ±5V R_S = 0V

FREQUENCY – kHz 32

48 64 80 96 112

16

Figure 37. Voltage Noise Density at +5 V from 0 Hz to 25 kHz

210

TEMPERATURE – 8C

SHORT-CIRCUIT CURRENT – mA

50

30

250

275250225 0 25 50 75 100 125 10

150 V_SY = +2.7V

240 230 220 0 20 40

I_SC2

I_SC+

Figure 40. Output Short-Circuit Current vs. Temperature

100

TEMPERATURE – 8C

OUTPUT VOLTAGE SWING – mV

250

200

0275250225 0 25 50 75 100 125 150

150 V_SY = +5.0V

25 50 75 125 175 225

R_L = 1kV

R_L = 10kV R_L = 100kV

Figure 43. Output Voltage to Supply Rail vs. Temperature

en – nV/ Hz

V_SY = +5V R_S = 0V

5 10

0

FREQUENCY – Hz 48

72 96 120 144 168

24

Figure 38. Voltage Noise Density at +5 V from 0 Hz to 10 Hz

220

TEMPERATURE – 8C

SHORT-CIRCUIT CURRENT – mA

100

60

2100

275250225 0 25 50 75 100 125 20

150 V_SY = +5.0V

280 260 240 0 40 80

I_SC2

I_SC+

Figure 41. Output Short-Circuit Current vs. Temperature

TEMPERATURE – 8C

POWER SUPPLY REJECTION – dB

150

145

125275250225 0 25 50 75 100 125 140

135

130

150 V_SY = +2.7V TO +5.5V

Figure 39. Power-Supply Rejection vs. Temperature

100

TEMPERATURE – 8C

OUTPUT VOLTAGE SWING – mV

250

200

0275250225 0 25 50 75 100 125 150

150 V_SY = +5.0V

25 50 75 125 175 225

R_L = 1kV

R_L = 10kV R_L = 100kV

Figure 42. Output Voltage to Supply Rail vs. Temperature

FUNCTIONAL DESCRIPTION

The AD855x family of amplifiers are high precision rail-to-rail operational amplifiers that can be run from a single supply volt- age. Their typical offset voltage of less than 1 µV allows these amplifiers to be easily configured for high gains without risk of excessive output voltage errors. The extremely small tempera- ture drift of 5 nV/°C ensures a minimum of offset voltage error over its entire temperature range of –40°C to +125°C, making the AD855x amplifiers ideal for a variety of sensitive measure- ment applications in harsh operating environments such as under-hood and braking/suspension systems in automobiles.

The AD855x family are CMOS amplifiers and achieve their high degree of precision through autozero stabilization. This autocorrection topology allows the AD855x to maintain its low offset voltage over a wide temperature range and over its operat- ing lifetime.

Amplifier Architecture

Each AD855x op amp consists of two amplifiers, a main amplifier and a secondary amplifier, used to correct the offset voltage of the main amplifier. Both consist of a rail-to-rail input stage, allowing the input common-mode voltage range to reach both supply rails.

The input stage consists of an NMOS differential pair operating concurrently with a parallel PMOS differential pair. The outputs from the differential input stages are combined in another gain stage whose output is used to drive a rail-to-rail output stage.

The wide voltage swing of the amplifier is achieved by using two output transistors in a common-source configuration. The output voltage range is limited by the drain to source resistance of these transistors. As the amplifier is required to source or sink more output current, the rDS of these transistors increases, raising the voltage drop across these transistors. Simply put, the output volt- age will not swing as close to the rail under heavy output current conditions as it will with light output current. This is a character- istic of all rail-to-rail output amplifiers. Figures 6 and 7 show how close the output voltage can get to the rails with a given output current. The output of the AD855x is short circuit protected to approximately 50 mA of current.

The AD855x amplifiers have exceptional gain, yielding greater than 120 dB of open-loop gain with a load of 2 kΩ. Because the output transistors are configured in a common-source configu- ration, the gain of the output stage, and thus the open-loop gain of the amplifier, is dependent on the load resistance. Open-loop gain will decrease with smaller load resistances. This is another characteristic of rail-to-rail output amplifiers.

Basic Autozero Amplifier Theory

Autocorrection amplifiers are not a new technology. Various IC implementations have been available for over 15 years and some improvements have been made over time. The AD855x design offers a number of significant performance improvements over older versions while attaining a very substantial reduction in de- vice cost. This section offers a simplified explanation of how the AD855x is able to offer extremely low offset voltages and high open-loop gains.

As noted in the previous section on amplifier architecture, each AD855x op amp contains two internal amplifiers. One is used as the primary amplifier, the other as an autocorrection, or nulling, amplifier. Each amplifier has an associated input offset voltage, which can be modeled as a dc voltage source in series with the noninverting input. In Figures 44 and 45 these are labeled as V_OSX, where x denotes the amplifier associated with the offset; A for the nulling amplifier, B for the primary amplifier. The open- loop gain for the +IN and –IN inputs of each amplifier is given as AX. Both amplifiers also have a third voltage input with an associated open-loop gain of B_X.

There are two modes of operation determined by the action of two sets of switches in the amplifier: An autozero phase and an amplification phase.

Autozero Phase

In this phase, all φA switches are closed and all φB switches are opened. Here, the nulling amplifier is taken out of the gain loop by shorting its two inputs together. Of course, there is a degree of offset voltage, shown as V_OSA, inherent in the nulling amplifier which maintains a potential difference between the +IN and –IN inputs. The nulling amplifier feedback loop is closed through φA₂ and VOSA appears at the output of the nulling amp and on CM1, an internal capacitor in the AD855x. Mathematically, we can ex- press this in the time domain as:

V_OA

[ ]

t ⁼A V_{A OSA}

[ ]

t ⁻B V_{A OA}

[ ]

t (1) which can be expressed as,

V t A V t

OA B

A OSA

[ ]

⁼ A

[ ]

+

1 (2) This shows us that the offset voltage of the nulling amplifier times a gain factor appears at the output of the nulling amplifier and thus on the CM1 capacitor.

V_IN+

V_IN2

V_OUT A_B

A_A FA

FB V_OSA

+

B_B C_M2

C_M1 FA FB

V_NB

V_NA 2B_A V_OA

Figure 44. Autozero Phase of the AD855x Amplification Phase

When the φB switches close and the φA switches open for the amplification phase, this offset voltage remains on CM1 and essentially corrects any error from the nulling amplifier. The voltage across CM1 is designated as VNA. Let us also designate V_IN as the potential difference between the two inputs to the primary amplifier, or VIN = (VIN+ – VIN–). Now the output of the nulling amplifier can be expressed as:

V_OA

[ ]

t ⁼A V_A

(

_IN

[ ]

t ⁻V_OSA

[ ]

t

)

⁻B V_{A NA}

^{[ ]}

t (3)

AD8551/AD8552/AD8554

V_IN+

V_IN2

V_OUT A_B

A_A FA

FB V_OSA

+

B_B C_M2

C_M1 FA FB

V_NB

V_NA 2B_A V_OA

Figure 45. Output Phase of the Amplifier

Because φA is now open and there is no place for CM1 to dis- charge, the voltage V_NA at the present time t is equal to the voltage at the output of the nulling amp VOA at the time when φA was closed. If we call the period of the autocorrection switching frequency TS, then the amplifier switches between phases every 0.5␣⫻␣ TS. Therefore, in the amplification phase:

V_NA

[ ]

t ⁼V_NA^_t⁻ T_S

 1

2 (4) And substituting Equation 4 and Equation 2 into Equation 3 yields:

V t A V t A V t

A B V t T

OA A IN A OSA B

A A OSA S

[ ]

⁼

[ ]

⁺

[ ]

⁻ A⁻



 

 +

1 2 1

(5)

For the sake of simplification, let us assume that the autocorrection frequency is much faster than any potential change in VOSA or V_OSB. This is a good assumption since changes in offset voltage are a function of temperature variation or long-term wear time, both of which are much slower than the auto-zero clock frequency of the AD855x. This effectively makes VOS time invariant and we can re- arrange Equation 5 and rewrite it as:

V t A V t A B V A B V

OA A IN B

A A OSA A A OSA

[ ]

⁼

[ ]

⁺

⁽

¹⁺

⁾

₁₊ A^{− (6)}

or,

V t A V t V

OA A IN OSAB

[ ]

^{= }_

[ ]

⁺₁₊ A^_ (7) We can already get a feel for the autozeroing in action. Note the VOS term is reduced by a 1 + BA factor. This shows how the nulling amplifier has greatly reduced its own offset voltage error even before correcting the primary amplifier. Now the primary amplifier output voltage is the voltage at the output of the AD855x amplifier. It is equal to:

V_OUT

[ ]

t ⁼A V_B

(

_IN

[ ]

t ⁺V_OSB

)

⁺B V_{B NB} (8) In the amplification phase, VOA = VNB, so this can be rewritten as:

Combining terms,

V t V t A A B A B V

B A V

OUT IN B A B A B OSA

A

B OSB

[ ]

⁼

[ ] (

⁺

)

⁺ ₁₊ ⁺ (10)

The AD855x architecture is optimized in such a way that A_A␣ =␣ A_B and B_A␣ =␣ B_B and B_A␣ >>␣ 1. Also, the gain product of AABB is much greater than AB. These allow Equation 10 to be simplified to:

V_OUT

[ ]

t ^≈V_IN

[ ]

t A B_{A A}⁺A V_A

(

_OSA⁺V_OSB

)

(11) Most obvious is the gain product of both the primary and nulling amplifiers. This AABA term is what gives the AD855x its extremely high open-loop gain. To understand how V_OSA and V_OSB relate to the overall effective input offset voltage of the complete amplifier, we should set up the generic amplifier equation of:

V_OUT = ×k

(

V_IN +V_OS^, _EFF

)

(12) Where k is the open-loop gain of an amplifier and V_{OS, EFF} is its effective offset voltage. Putting Equation 12 into the form of Equation 11 gives us:

V_OUT

[ ]

t ^≈V_IN

[ ]

t A B_{A A}⁺V_OS^, _EFFA B_{A A} (13) And from here, it is easy to see that:

V V V

OS EFF OSAB OSB A

, ≈ +

(14)

Thus, the offset voltages of both the primary and nulling ampli- fiers are reduced by the gain factor B_A. This takes a typical input offset voltage from several millivolts down to an effective input offset voltage of submicrovolts. This autocorrection scheme is what makes the AD855x family of amplifiers among the most precise amplifiers in the world.

High Gain, CMRR, PSRR

Common-mode and power supply rejection are indications of the amount of offset voltage an amplifier has as a result of a change in its input common-mode or power supply voltages. As shown in the previous section, the autocorrection architecture of the AD855x allows it to quite effectively minimize offset volt- ages. The technique also corrects for offset errors caused by common-mode voltage swings and power supply variations.

This results in superb CMRR and PSRR figures in excess of 130 dB. Because the autocorrection occurs continuously, these figures can be maintained across the device’s entire temperature range, from –40°C to +125°C.

Maximizing Performance Through Proper Layout

To achieve the maximum performance of the extremely high input impedance and low offset voltage of the AD855x, care should be taken in the circuit board layout. The PC board sur- face must remain clean and free of moisture to avoid leakage currents between adjacent traces. Surface coating of the circuit board will reduce surface moisture and provide a humidity barrier, reducing parasitic resistance on the board. The use of